This invention relates to a method and apparatus for detecting signals. The invention is especially, but not exclusively, applicable to systems which may be subject to an unknown Doppler frequency shift affecting modulated coherent signals used for communication and ranging purposes, and particularly substantially continuous signals modulated by random or chaotic waveforms utilized by obstacle detection or collision avoidance systems.
Such systems are designed to operate in multiuser and often hostile environments, and are intended for a wide range of applications, including automotive applications, industrial robotics and unmanned vehicle navigation.
FIG. 1 is a block diagram of a typical microwave obstacle-detection system. The system comprises a signal generator 1 that produces a substantially continuous waveform x(t) with suitable bandwidth to provide required range resolution. The waveform x(t) may be deterministic (periodic or aperiodic), chaotic or purely random.
The system also has a microwave oscillator 2 that generates a sinusoidal signal with required carrier frequency, a modulator 3 that modulates one or more of the parameters (such as amplitude, phase, or frequency) of the carrier signal with the modulating waveform x(t), a power amplifier (PA) 4 that amplifies the modulated carrier signal to a required level, a microwave transmit antenna (TA) 5 that radiates an electromagnetic wave representing the modulated carrier signal towards an obstacle 6, a microwave receive antenna (RA) 7 that receives an electromagnetic wave reflected back by the obstacle 6, an input amplifier (IA) 8 that amplifies a signal provided by the receive antenna (RA) 7, and a coherent demodulator 9 that processes jointly the reference carrier signal supplied by the oscillator 2 and a signal supplied by the input amplifier (IA) 8 to reconstruct a time-delayed replica y(t) of the modulating waveform x(t).
The modulating waveform x(t) and its time-delayed replica y(t) are then processed jointly during a specified time interval by a suitable processor 10, such as correlator, to produce an estimate of the unknown time delay that is proportional to the distance (range) between the system and the obstacle 6.
FIG. 2 shows an example of a correlation function of a synchronous random binary waveform.
When there occurs a relative movement between a ranging system and an obstacle of interest, an electromagnetic wave reflected back by the obstacle and received by a coherent system will exhibit a Doppler frequency shift. The value xcfx89D0 of this (angular) frequency shift can be determined from:       ω    D0    =                    2        ⁢                  υ          0                    c        ⁢          ω      0      
where xcexd0 is the radial speed (i.e., the range rate) of a relative movement between the system and the obstacle and xcfx890 is the (angular) carrier frequency of a transmitted electromagnetic wave having velocity c.
The signal reflected back by a moving obstacle can be expressed as:
zr(t)=xcex3x(txe2x88x92xcfx840)cos [(xcfx890+xcfx89D0)(txe2x88x92xcfx840)+xcex8]
where xcex3 is the round-trip attenuation, xcfx840 is the delay corresponding to the range D, xcfx89D0 is the Doppler frequency shift, and xcex8 is an unknown constant phase shift. Strictly speaking, the value of xcfx840 cannot be constant for a nonzero Doppler frequency xcfx89D0. However, in most practical cases it is assumed that the small changes in range D, of the order of the wavelength of the carrier frequency, cannot be discerned when determining a short-time time-delay estimate. Such assumption justifies decoupling delay and Doppler frequency measurements.
A baseband signal corresponding to the received signal zr(t) has the form
y(t)=x(txe2x88x92xcfx840)cos(xcfx89D0t+xcfx86)
where xcfx86 is an unknown phase shift.
The correlator performs the following operation             R              x        ⁢                  xe2x80x83                ⁢        y              ⁢          (      τ      )        =            1              T        0              ⁢                  ∫        0                  T          0                    ⁢                        x          ⁢                      (                          t              -              τ                        )                          ⁢                  x          ⁢                      (                          t              -                              τ                0                                      )                          ⁢                  cos          ⁢                      (                                                            ω                  D0                                ⁢                t                            +              ϕ                        )                          ⁢                  ⅆ          t                    
where the integral is evaluated for a plurality of hypothesized time delays xcfx84min less than xcfx84 less than xcfx84max. When the observation interval T0 is much shorter than one period (2xcfx80/xcfx89D0) of the Doppler frequency, the value of cos(xcfx89D0tm+xcfx86) is almost constant during T0, so that the correlation integral can be approximated by             R              x        ⁢                  xe2x80x83                ⁢        y              ⁢          (      τ      )        ≈            1              T        0              ⁢          cos      ⁢              (                                            ω              DO                        ⁢                          t              m                                +          ϕ                )              ⁢                  ∫        0                  T          0                    ⁢                        x          ⁢                      (                          t              -              τ                        )                          ⁢                  x          ⁢                      (                          t              -                              τ                0                                      )                          ⁢                  ⅆ          t                    
where the time instant tm is taken in the middle of the observation interval T0.
When the correlation integral is calculated repeatedly for successive short processing intervals, each of duration T0, the sequence of observed correlation functions may be represented by the plot of FIG. 3. The rate of change (in time) of the correlation function Rxy(xcfx84) will correspond to the Doppler frequency xcfx89D0. The value of this frequency can be determined by applying some suitable form of spectral analysis.
FIG. 4 is a block diagram of a conventional correlator, comprising variable delay line 11, multiplier 12 and integrator 13, followed by a spectrum analyser 14. When the total number of successive short processing intervals is large enough, the frequency spectrum S(xcfx89), observed at the output of the analyser, will exhibit a pronounced peak at the Doppler frequency xcfx89D0.
FIG. 5 is a block diagram of a multichannel correlator that uses a tapped delay line 15 to cover the entire interval of hypothesized time delays, xcfx84min less than xcfx84 less than xcfx84max, in J steps of unit delay xcex94 using delay circuits 16. Multipliers 17 process the delayed signals wiith y(t) and provide the outputs to integrators 18. The required spectral analysis may be performed by a digital processor 19 implementing the Discrete Fourier Transform (DFT). The principle of operation is thus similar to that of FIG. 4.
The systems shown in FIGS. 4 and 5, and also other similar known systems, attempt to approximate the correlation integral with two parameters of interest, delay time xcfx84 and Doppler frequency xcfx89D by combining a time sequence of correlation functions determined over a number of relatively short observation intervals T0. Because of this approximation, such systems can only provide suboptimal solutions to the problem of joint estimation of time delay and Doppler frequency.
Another prior art technique involves determining values of the following correlation integral             R              x        ⁢                  xe2x80x83                ⁢        y              ⁢          (              τ        ,                              ω            D                    ;          ϕ                    )        =            1              T        0              ⁢                  ∫        0                  T          0                    ⁢                        x          ⁢                      (                          t              -              τ                        )                          ⁢                  cos          ⁢                      (                                                            ω                  D                                ⁢                t                            +              ϕ                        )                          ⁢                  y          ⁢                      (            t            )                          ⁢                  ⅆ          t                    
for the entire interval of hypothesized time delays
xcfx84min less than xcfx84 less than xcfx84max,
and also for the entire interval of hypothesized Doppler frequencies
xcfx89Dmin less than xcfx89D less than xcfx89Dmax;
the unknown phase xcfx86 should also be varied over the interval (0, 2xcfx80). In this case, values of the correlation integral, determined for some prescribed range of argument values (xcfx84,xcfx89D), define a two-dimensional correlation function. The specific values of arguments xcfx84 and xcfx89D, say xcfx840 and xcfx89D0, that maximise the two-dimensional correlation function provide estimates of unknown time delay and unknown Doppler frequency.
FIG. 6 is a block diagram of a suitable system capable of determining values of the correlation integral for some specified time delays xcfx84 and Doppler frequencies xcfx89D. The system is formed by a correlator like that of FIG. 4, including variable delay line 11, multiplier 12 and integrator 13. However, the signal x(t) is first applied to a multiplier 20 which multiplies it by a variable Doppler signal cos(xcfx89Dt+xcfx86).
The procedure of adjusting the unknown value of phase xcfx86 can be avoided by suitably combining the values of the following two integrals:                                           R                          x              ⁢                              xe2x80x83                            ⁢              y              ⁢                              xe2x80x83                            ⁢              c                                ⁡                      (                          τ              ,                                                ω                  D                                ;                ϕ                                      )                          =                              1                          T              0                                ⁢                                    ∫              0                              T                0                                      ⁢                                          y                ⁡                                  (                  t                  )                                            ⁢                              x                ⁡                                  (                                      t                    -                    τ                                    )                                            ⁢                              cos                ⁡                                  (                                                            ω                      D                                        ⁢                    t                                    )                                            ⁢                              ⅆ                t                                                                        (        1        )                                                      R                          x              ⁢                              xe2x80x83                            ⁢              y              ⁢                              xe2x80x83                            ⁢              s                                ⁡                      (                          τ              ,                                                ω                  D                                ;                ϕ                                      )                          =                              1                          T              0                                ⁢                                    ∫              0                              T                0                                      ⁢                                          y                ⁡                                  (                  t                  )                                            ⁢                              x                ⁡                                  (                                      t                    -                    τ                                    )                                            ⁢                              sin                ⁡                                  (                                                            ω                      D                                        ⁢                    t                                    )                                            ⁢                              ⅆ                t                                                                        (        2        )            
evaluated for the entire interval of hypothesized time delays
xe2x80x83xcfx84min less than xcfx84 less than xcfx84max
and the entire interval of hypothesized Doppler frequencies
xcfx89Dmin less than xcfx89D less than xcfx89Dmax.
In many industrial and automotive applications, systems capable of detecting moving obstacles and employing substantially continuous modulated microwave (or other coherent radiation), developed in accordance with the prior art, will be too complicated and also too expensive.
Furthermore, obstacle-detection systems designed for multiuser environments will preferably exploit some suitable form of random or chaotic modulation which may preclude the use of conventional Doppler signal processors constructed in accordance with the prior art.
Aspects of the present invention are set out in the accompanying claims.
In accordance with a further aspect of the invention, representations of a transmitted signal and a received signal are processed together in such a manner as to determine whether or not a target signal is present (this target signal possibly representing an object reflecting the transmitted signal). An auxiliary representation, for example formed of one or more auxiliary signals, is introduced into the process. The auxiliary representation includes frequencies distributed throughout a range which corresponds to anticipated frequency modifications (for example Doppler shifts) in the target signal, as a result of which the system has response characteristics such that a target signal exhibiting frequency modifications within this range will give rise to a significant system output, whereas frequency modifications outside this range will not.
Such an arrangement avoids the need for determining the value of the frequency modification (e.g. Doppler shift) but enables the rapid detection of any target signal incorporating a frequency modification of interest, using a simple structure. The above-mentioned arrangement is thus particularly useful in applications where it is sufficient to discriminate between stationary obstacles and obstacles moving with specified velocities, and for applications in which obstacles are placed into a number of classes according to obstacle velocity, for example zero/low, moderate/average, high/very high.
Preferred aspects of the present invention are directed to the nature of the auxiliary representation and give rise to a number of advantages as explained below.
Preferably, the auxiliary representation is in the form of finite-duration signal portions comprising multiple frequencies within the range of interest, and preferably distributed throughout this range. The duration of the signal portions is selected so as to maintain orthogonality of the components of the respective frequencies. By using finite-duration portions and selecting the shapes of these portions appropriately, it is possible to extend the system response characteristics such that a significant output is produced for not only target signal frequencies equal to those of the signal components used to construct the auxiliary representation, but also adjacent frequencies.
Preferably, each signal portion is created using a window function which results in a significant frequency response for all frequencies within the range of interest. However, it is difficult to arrange for a signal portion to produce a flat response throughout the frequency range of interest.
For this reason, in accordance with a preferred embodiment of the invention, the auxiliary representation comprises different finite-duration signal portions, each comprising components having a respective set of frequencies, and arranged such that they give rise to complementary frequency response characteristics. Accordingly, by combining the results achieved using the respective different auxiliary signal portions, it is possible to get an overall frequency response characteristic which is significantly flatter than if only one type of signal portion were used.
Preferably, each signal portion comprises components of frequencies interleaved with those of the other signal portion or portions.
An indicated by FIG. 3, at any given instant the correlation function may have a low level depending upon the phase of the Doppler shift. In order to ensure a significant output irrespective of the unknown phase, the auxiliary representation preferably comprises a first representation and a second representation, the components of respective frequencies in the first representation having a quadrature phase relationship with the corresponding frequency components in the second representation. By combining the responses achieved as a result of theses two auxiliary signals, the output is substantially unaffected by the phase of the Doppler shift.
Accordingly, in the preferred embodiment, there are at least four types of auxiliary signal portions: signal portions which differ in that they comprise frequencies interleaved with frequencies of other portions, and signal portions which differ in that their frequency components are in quadrature relationship with those of other portions.
The preferred embodiment uses four types of signal portions, but it would be possible to produce a system which uses more types, while maintaining the advantages indicated above. For example, the frequency components may be split into three or more sub-sets.
It would be desirable to synthesise digitally the auxiliary representation. In order to facilitate this, the auxiliary representation preferably exhibits small values of the peak factor, i.e., the peak value divided by its root-mean square value. Minimising this enables the waveform to be represented digitally with a minimum number of bits for a specified reconstruction error.
A small peak factor can be achieved by appropriate selection of the initial phase values of the signal components for each signal portion.
In the preferred embodiment described below, the transmitted and received representations are signals which are combined by correlation. However, other arrangements are possible. For example, the received signal may be modified (e.g. multiplied) by the auxiliary representation and then delivered to a filter matched to the characteristics of the transmitted signal. This use of matched filters, which utilise a representation of the transmitted signal rather than the signal itself, is, per se, known in the art.
In another alternative embodiment, the correlator is replaced by a suitably modified version of the time-delay discriminator described in WO-A-00/39643, the contents of which are incorporated herein.
Although the present invention is being described primarily in the context of detection of obstacles which produce a Doppler shift in a reflected signal, the invention is applicable also in other areas. For example, the invention is applicable to communication systems in which the local oscillators in the transmitter and receiver may not be accurately matched; a frequency offset between the oscillators would give rise to a frequency shift similar to a Doppler shift. The invention thus permits use of equipment which has wider tolerances, and is thus less expensive, than prior art arrangements.
The invention relates both to a method of signal processing and to apparatus operating according to such a method.
Arrangements embodying the invention will now be described by way of example with reference to the accompanying drawings.